System and method for applying multi-tone ofdm based communications within a prescribed frequency range

ABSTRACT

According to one embodiment of the invention, an integrated circuit comprises an encoding module, a modulation module and a spectral shaped module. The encoding module includes an interleaver that adapted to operate in a plurality of modes including a first mode and a second mode. The interleaver performs repetitive encoding when placed in the second mode. The modulation module is adapted to compensate for attenuations that are to be realized during propagation of a transmitted signal over the power line. The spectral shaped module is adapted to compensate for amplitude distortion and further compensates for attenuations that will be realized during propagation of the transmitted signal over the power line.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.13/680,259 (U.S. Pat. No. 8,780,691), filed Nov. 19, 2012, which is acontinuation of U.S. patent application Ser. No. 12/478,618 (U.S. Pat.No. 8,315,152), filed Jun. 4, 2009. This application claims the benefitof U.S. Provisional Application No. 61/059,684, filed on Jun. 6, 2008.The disclosures of the above applications are incorporated herein byreference in its their entirety.

FIELD

Embodiments of the invention generally relate to a field ofcommunications over a power line. More specifically, one embodiment ofthe invention relates to a system and method for applying multi-tone,orthogonal frequency division multiplexing (OFDM) based communicationsover a prescribed frequency region.

BACKGROUND

For many years, both direct current (DC) and alternating current (AC)power lines have been used in order to transfer data from one device toanother. Recently, there has been a growing need for new datatransmission services and applications that are more reliable andsupport higher data rates over these power lines. For instance, remotemetering, smart grids, industrial and home automation are merely some ofthe upcoming applications that are currently using power lines tosupport data communications and greater use of these power lines isexpected.

One primary disadvantage in using power lines for data transfer is thatpower lines are hostile environments. In fact, communication channelssupported by these AC power lines tend to experience severe non-linearbehavior. In particular, channel characteristics and parameters may varydue to changes in frequency, location, time and even the type ofequipment deployed. As an example, the impedance of a power line mayappear to be 1-2 ohms (Ω), but as the frequency of signaling applied tothe power line increases, the impedance of the power line alsoincreases. This increased impedance causes increased signal noise thatmay hamper proper detection of the data at an intended destination.

FIG. 1 illustrates an exemplary noise power distribution 100 withfrequency bands 110-112 supported by European (CELENEC), United States(FCC) and Japan (ARIB) power line standards respectively, as outlined inTable A.

TABLE A LOW HIGH FREQ FREQ STANDARDS (KHz) (KHz) FCC 10 480 ARIB 10 450CELENEC A 9 95 CELENEC B 95 125 CELENEC C 125 140 CELENEC B, C 95 140

As illustrated in FIG. 1, a low frequency region from three kilohertz (3kHz) to 500 kHz is especially susceptible to interference such asnarrowband interference and/or intersymbol interference (ISI), which mayoccur if orthogonal frequency division multiplexing (OFDM) is used asthe selected data transmission scheme.

SUMMARY

OFDM is a multi-carrier modulation scheme that subdivides the availablefrequency spectrum into a number of narrowband channels (e.g., around100 channels). The carriers for each channel may be spaced much closertogether than Frequency Division Multiplexing (FDM) for example, becauseeach carrier is configured to be orthogonal to its adjacent carriers.This orthogonal relationship may be achieved by setting each carrier tohave an integer number of cycles over a symbol period. Hence,theoretically, there is no interference between the carriers themselves,albeit interference caused by environmental conditions may exist.

Besides interference, the low frequency region is highly susceptible toimpulsive noise and group delay. “Impulsive noise” is characterized as ashort peak pulse substantially less than one second (e.g., a fewmicroseconds) and with a sharp rise time above the continuous noiselevel experienced by the signal. “Group delay” is a measure of the rateof change of the phase with respect to frequency. As an effect ofnon-constant group delay, phase distortion will occur and may interferewith accurate data recovery.

Prior power line communications have virtually avoided using OFDM-basedcommunication techniques within the lower frequency region. One reasonfor such avoidance can be ascertain by review of the noise powerdistribution of FIG. 1. Between 10 kHz and 148 kHz (Celenec bands shownas region A 110), the effects of noise on an input signal of a givenfrequency level is about 10-30 decibels (dB) stronger than the noiseexperienced by signals at approximately 150 kHz as represented in regionB 120. Moreover, the channel frequency response varies drasticallywithin the region between 10 kHz-150 kHz, resulting in severe amplitudeand phase distortion.

BRIEF DESCRIPTION OF THE DRAWINGS

Features and advantages of embodiments of the invention will becomeapparent from the following detailed description in which:

FIG. 1 is an exemplary depiction of noise power distribution over apower line.

FIGS. 2A and 2B are exemplary block diagrams of a networked deviceutilizing power lines for data transfer.

FIG. 3 is an exemplary block diagram of an integrated circuit thatcontrols communications over a power line of FIG. 2A and/or FIG. 2B.

FIG. 4 illustrates an exemplary embodiment of transmitter/receivercombination of the networked device as described in FIG. 2A and/or FIG.2B.

FIGS. 5A-5H illustrate more detailed embodiments of a transmitter of thenetworked device.

FIG. 6 illustrates a more detailed embodiment of a receiver of thenetworked device.

FIG. 7 illustrates the operations of a frequency-domain pre-emphasisfilter implemented within the transmitter of the networked device ofFIG. 5A.

DETAILED DESCRIPTION

Embodiments of the invention set forth in the following detaileddescription generally relate to methods, apparatus, software, andsystems for applying multi-tone orthogonal frequency divisionmultiplexing (OFDM) based communications within a prescribed frequencyband such as between 10 kHz and 600 kHz. According to one embodiment ofthe invention, the prescribed frequency band may involve frequenciesless than 500 kHz and is programmable. For instance, the prescribedfrequency band may be within 3 kHz-148 kHz (Celenec, European standard);10 kHz-480 kHz (FCC, United States standard); 10 kHz-450 kHz (ARIB,Japanese standard) or the like. Of course, the invention may beapplicable to chosen frequency bands greater than 500 kHz.

Of course, it is contemplated that a combination of a feedback analogautomatic gain control (AGC) followed by a digital AGC, tone mappingmodule and adaptive notching module (described below) may be adapted tohandle communications through a HV/LV transformer.

In the following description, certain terminology is used to describecertain features of the invention. For example, the terms “logic” and“module” broadly represent hardware, software, firmware or anycombination thereof. The term “power line” generally represents a mediumadapted to carry direct current (DC) or alternating current (AC).

Referring now to FIG. 2A, a general embodiment of a network system 200adapted to support multi-tone OFDM base communications is shown. Herein,a networked device 210 is connected to a power line 220 via a power cord230. In general, “networked device” 210 is any equipment that is capableof transmitting and/or receiving information over power line 220.Examples of different types of networked devices include, but are notlimited or restricted to a computer, a router, an access point (AP), awireless meter, a networked appliance, an adapter, or any devicesupporting capability to a wired or wireless network.

According to one embodiment of the invention, networked device 200includes Physical Layer (PHY) logic 240 that is adapted to process datafor transmission over power cord 230 to power line 220. PHY logic 240comprises circuitry and perhaps software to support electrical andmechanical functionality with power line 220, where these electrical ormechanical connections provide an interface to format signals fortransmission or receipt over power lines 220. Such functionality mayinclude, but is not limited or restricted to digital-to-analogconversion, analog-to-digital conversion, modulation, and/or errorcorrection. This functionality is described in further detail withrespect to the receiver and transmitter modules set forth in FIGS. 4-6.

According to one embodiment of the invention, PHY circuitry 240 mayinclude circuitry, such as an OFDM Power Line Communication PhysicalLayer (OFDM PLC PHY) circuitry 310 of FIG. 3, which enables networkconnectivity over power lines 220 via power cord 230. Power cord 230includes a two-prong, or three-prong plug that is placed into a wallsocket 250, which is connected to an end of power line 220.

It is contemplated that networked device 200 may be configured as anadapter 260 in communication with a device (e.g., computer, AP, etc.)that is to be connected to the power line network but does not featurethe necessary PHY circuitry. As shown in FIG. 2B, adapter 260 isimplemented with OFDM PLC PHY circuitry 310 and operates as anintermediary between the device and power line 220. For thisimplementation, it is contemplated that adapter 260 includes a connector270 that enables communications with the device and a power cord 280that establishes a connection with power line 220. Connector 270 mayinclude any serial and/or parallel port such as, for example, a RJ-11jack, Universal Serial Bus (USB) port, or the like.

Referring now to FIG. 3, an exemplary embodiment of an integratedcircuit that controls communications over power line 220 of FIGS. 2A and2B is shown. Herein, a combination of physical (PHY) and media accesscontrol (MAC) layers is implemented within a single integrated circuit300. Herein, the PHY layer, namely OFDM PLC PHY circuitry 310, uses OFDMtechnology with DBPSK modulation and forward error correction (FEC) toprovide robust communication in the presence of narrowband interference,group delay, jammer signals, impulsive noise, and frequency-selectiveattenuations. OFDM PLC PHY circuitry 310 includes a transmitter and areceiver as described below.

As shown, OFDM PLC PHY circuitry 310 is controlled by a microcontroller320 (e.g., 16-bit RISC MAXQ® microcontroller) that is placed on-chip aswell. Moreover, integrated circuit 300 further includes flash memory330, static random access memory 340, and serial interfaces 350 (e.g.,universal asynchronous receiver/transmitter “UART”, System PacketInterface “SPI”, Inter-Integrated Circuit “I²C”) for communication amongdevices on the power line network.

Referring to FIG. 4, an exemplary embodiment of a transmitter 400integrated within OFDM PHY circuitry 310 and adapted for communicationswith an OFDM receiver is shown. Herein, according to this embodiment ofthe invention, transmitter 400 comprises a plurality of modulesincluding an encoding module 410, a modulation module 420, and aspectral shaping module 430.

As shown in FIGS. 5A and 5B, an exemplary embodiment of transmitter 400,especially encoding module 410, is shown. In general, a functionality ofencoding module 410 is to randomize the incoming data and to inserterror correction data into the transmitted signal. Herein, for thisembodiment of the invention, encoding module 410 comprises a scrambler500, a Reed-Solomon encoder 510, a convolutional encoder 520 and aninterleaver sub-system 530 that may operate in either Normal mode orRobust (ROBO) mode, where interleaver sub-system 530 performs repetitiveencoding when operating in ROBO mode. In other words, ROBO mode providesextra redundancy to facilitate communications in frequency selective,severely attenuated channels. Optional padding may be performed byencoding module 410.

Transmitter 400 is adapted to receive its input bits in a packet fromthe Media Access (MAC) Layer. Encoding module 410 may add parity bits tothe data and the packet increases in size as it passes through encodingmodule 410. At the end of encoding module 410, the final packet may besegmented into small packet(s) by “Un-Buffer” block 540 of FIG. 5B sothat each small packet may be fitted into one OFDM symbol. The size ofthis small packet depends on the number of carriers used in each OFDMsymbol.

For example, in FCC band, the packet size becomes equal to 100 bits. Inorder to understand the size of data as well as signal dimensions ateach various points in the transmitter 400, the number of bits carriedby a packet (e.g., PHY frame) may be obtained as set forth by equation(1):

N _(F) =N _(G) =Ncarr×Ns  (1)

Herein, “N_(F)” and “N_(G)” represent the size of packet (signal) atpoints (F) and (G) of FIG. 5B, respectively. Where “Ncarr” is the numberof carriers in each OFDM symbol and “Ns” is the number of symbols perPHY frame. Note that the interleaver does not change the size of packet.The number of bits at point (E) may be computed by equation (2):

N _(E) =N _(F) ×R  (2)

Herein, the value “R” may be one “1” for Normal mode and a fractionbased on repetition level (e.g., “¼”) for ROBO Mode. In order todetermine M, the number of zeros that may be needed to pad at the outputof convolutional encoder 520, the maximum number of Reed-Solomon (RS)bytes needs to be computed. The maximum number of RS bytes (MaxRSbytes)at the output of RS encoder 510 can be obtained as shown in equation(3):

MaxRSbytes=floor((N _(E×CCRate−CCZeroTail)/)8)  (3)

Where “CCRate” and “CCZeroTail” are the convolutional code rate (½) andthe number of zeros to be added to the input of convolutional encoder520 (to terminate the states to zero state), respectively. Thedenominator “8” refers to the length of each RS word that is one byte.Therefore, the value of “M” may be obtained by (see Table 1 below):

M=N _(E)-((MaxRSbytes×8)+6)×2  (4)

And the number of bits at point (D), (C) and (B) now may be calculatedby:

N _(D) =N _(E) −M

N _(C) =N _(D)/2

N _(B) =N _(C)−6

TABLE 1 # of zeroes padded after convolutional encoder ROBO (bits)Normal (bits) FCC M = 12 M = 4 ARIB M = 6 M = 12 CELENEC A M = 12 M = 4CELENEC B M = 4 M = 4 CELENEC C M = 4 M = 4 CELENEC B, C M = 4 M = 4

Finally, considering the fact the number of parity bytes in RS code maybe equal to 8, the packet size delivered by MAC to the physical layermay be given by:

N _(A)=(N _(B)−8)×8

The input packet to the physical (PHY) layer for various band and bothnormal and ROBO modes may be summarized in Table 2 below.

TABLE 2 Packet size delivered by MAC layer to PHY layer ROBO (bits)Normal (bits) FCC 424 1928 ARIB 392 1784 CELENEC A 304 1448 CELENEC B168 888 CELENEC C 168 888 CELENEC BC 288 1368

Referring still to FIGS. 5A and 5B, scrambler 500 randomly distributesthe incoming data. For instance, according to one embodiment of theinvention, incoming data 501 may undergo an exclusive-OR (XOR) operationwith a repeating pseudo-random sequence to produced scrambled data 502as shown in FIG. 5C. Herein, according to one embodiment of theinvention, a generator polynomial S(x)=x⁷+x⁴+1 is used where the bits inscrambler 500 are initialized to all ones at the start of processingeach PHY frame.

Thereafter, scrambled data 502 is routed to a first error correctionscheme, namely a Reed-Solomon (RS) encoder 510. Read-Solomon encoder 510is responsible for recovering data destroyed by impulsive noise. Hence,RS encoder 510 recovers data associated with a tone experiencingimpulsive noise and outputs such data 511 to convolutional encoder 520.

Herein, according to one embodiment of the invention, data fromscrambler 500 may be encoded by RS encoder 510 created by shortening theRS code (255,247, t=4) and (255,239, t=8). The “RS symbol word length,”namely the size of the data words used in the Reed-Solomon block, may befixed at 8-bits. The value of t (number of word errors that can becorrected) can be either 4 or 8 for different standards. For CENELEC B&Cand ROBO, the RS parity of 8-bytes (corresponding to t=4) should beused. The number of parity words in a RS-block is thus “2t” words.

The number of non-parity data words (bytes) in Reed-Solomon block may beprovided in Table 3 below. The first bit in time from scrambler 500 maybecome the most significant bit “MSB” of that symbol. Each RS encoderinput block (e.g., 247 symbols) is conceptually formed by one or morefill symbols (“00000000”) followed by the message symbols. Output of RSencoder 510 (with fill symbols discarded) may proceed in time from firstmessage symbol to last message symbol followed by parity symbols, witheach symbol shifted out most significant bit first.

Code Generator Polynomial g(x)=(x−α ¹)(x−α ²)(x−α ³) . . . (x−α ⁸)

Field Generator Polynomial: p(x)=x ⁸ +x ⁴ +x ³ +x ²+1(435 octal)

TABLE 3 RS encoder input/output packet size Normal Mode ROBO ModeN_(A)/N_(B) N_(A)/N_(B) (Bytes) (Bytes) FCC 241/249 53/61 ARIB 223/23149/57 CENELEC A 181/189 38/46 CENELEC BC 171/179 36/44 CENELEC B 111/11921/29 CENELEC C 111/119 21/29

The representation of α⁰ is “00000001”, where the left most bit of thisRS symbol is the MSB and is first in time from scrambler 500 and is thefirst in time out of RS encoder 510.

Next, convolutional encoder 520 is responsible for recovering thoseportions of data 511 that are affected by white noise present withindata 511 due to Gaussian noise from the environment. This white noisecauses degradation of the data associated with each tone. While thedegradation of the data cannot be eliminated, its effect can bemitigated through forward error correction (FEC) techniques that areperformed by convolutional encoder 520.

Herein, according to one embodiment of the invention, the bit stream atthe output of the Reed-Solomon block may be encoded with a standardrate=1/2, K=7 convolutional encoder 520. The tap connections are definedas x=0b1111001 and y=0b1011011, as shown in FIG. 5D.

When the last bit of data to convolutional encoder 520 has beenreceived, convolutional encoder 520 inserts six tail bits, which may berequired to return convolutional encoder 520 to the “zero state”. Thismay improve the error probability of the convolutional decoder, whichrelies on future bits when decoding. The tail bits may be defined as sixzeros. The number of bits at the input and the output of convolutionalencoder 520 may be given in Table 4.

TABLE 4 Convolutional encoder input/output packet sizes Normal Mode ROBOMode N_(A)/N_(B) N_(A)/N_(B) (bits) (bits) FCC 1998/3996 494/988 ARIB1854/3708 462/924 CENELEC A 1518/3036 374/748 CENELEC BC 1438/2876358/716 CENELEC B  958/1916 238/476 CENELEC C  958/1916 238/476

Next, interleaver sub-system 530 is used to intelligently randomize thedata in frequency (subcarriers) and time in order to achieve frequencyand time diversity. In other words, data may be repositioned indifferent frequency channels and even in different symbols (OFDM).Interleaver sub-system 530 generally is adapted to operate in twodifferent modes: Normal mode 532 and Robust (or ROBO) mode 534. InNormal mode, interleaver sub-system 530 randomizes the data in frequencyand time. In ROBO mode, however, interleaver sub-system 530 not onlyrandomizes the data in frequency and time, but also facilitatescommunications over a degraded channel by reducing the data rate andperforming repetition coding.

This repetition coding involves the operation of repeating the data bitsfrom convolutional encoder 520 multiple times (e.g., four times) andstatistically placing these bits at different frequencies and withindifferent symbols in order to increase reliability. Whether interleaversub-system 530 operates in Normal mode or ROBO mode is controlled bymicrocontroller 320 of FIG. 3 based on signaling from a correspondingreceiver that is currently in communications with transmitter 400. Suchsignaling is the result of the channel estimation module of the receivermeasuring the SNR of the channel and using this information to decidethe mode of operation. Thereafter, the receiver communicates thisinformation for receipt by transmitter 400.

Herein, according to one embodiment of the invention, an interleaver 535being part of interleaver sub-system 530 may be used for both Normal andROBO modes. As shown in FIG. 5E, interleaver 535 comprises foursub-interleavers 590, 591, 592 and 593. All sub-interleavers 590-593 maybe the same and have same architectures. A flipping block 594 precedessub-interleavers 591 and 592. This means that packets provided alongthese two branches may be written backward from the end of their inputsto the beginning. The segmentation of input packet 595 into a pluralityof sub-packets that are transmitted along each branch is performed by ademultiplexer 596. Demultiplexer 596 may be different for each mode(Normal and ROBO modes) as described below.

A. Demultiplexer for Normal Mode

The input data bits may be applied to each branch alternatively. Thismeans that the first data bit may be applied to first sub-interleaver590; the second bit may be applied to second sub-interleaver 591, and soon.

B. Demultiplexer for Robo Mode

As shown in FIG. 5F, the input packet may be repeated four times (herebythe repeat code (by 4) is set). However, the data for the third andfourth branch 592 and 593 may be read from bit number 233 (assuming thefirst number is 1) and the reading is wrapped around at the end of thepacket. This means that when the end of packet is reached, it may startthe reading from the beginning and stops at bit number 232. This methodis graphically illustrated in FIG. 5F.

C. Helical Scan Interleaver

The helical scan interleaver (sub-interleaver 590-593) may write theinput bits row-by-row in a matrix and read them diagonal wise. Thenumber of rows in each sub-interleaver 590-593 may be equal to 10 andthe number of columns may be computed as shown below and illustrated inTable 5:

Number of Columns=Number of symbols*Number of carries/40

TABLE 5 # interleaver rows and columns for bands No. of Rows No. ofColumns FCC 10 100 ARIB 10 93 CENELEC A 10 76 CENELEC B 10 48 CENELEC C10 48 CENELEC BC 10 72

For illustrative purposes, as shown in FIG. 5G, for the input array [1,2, 3, 4, 5, 6, 7, 8, 9, 10, 11, 12, 13, 14, 15, 16, 17, 18, 19, 20, 21,22, 23, 24, 25, 26, 27, 28, 29, 30, 31, 32] where the data is writtenrow by row. Reading diagonal wise, starting from index position (0,0),going through traces 597, 598, 599, the output array may be given by:[1, 10, 19, 28, 5, 14, 23, 32, 9, 18, 27, 4, 13, 22, 31, 8, 17, 26, 3,12, 21, 30, 7, 16, 25, 2, 11, 20, 29, 6, 15, 24].

Note that the input packet to the second and third sub-interleavers 591and 592 of FIG. 5E may be flipped. In other words, the packet may bewritten from the end of array. The outputs of sub-interleavers 590-593may be finally concatenated by logic 597, which means that the wholeoutput packet of the first interleaver 590 may be sent first, the wholeoutput packet of the second interleaver 591 may be sent second, and soon so forth (e.g., bit by bit alternation may not be applied to theoutput). An exemplary C code to implement the interleaver is givenbelow:

R = 10; //Number of rows C = Ns * Ncarr / 40; //Number of columns i=0;//writing into the memory row by row for (n=0;n<R;n++) { for(m=0;m<C;m++) { a[n][m]= x[i]; // x is the input array i = i + 1; } }//reading the memory diagonal wise i=0; for (n=0;n<R;n++) { k = n; for(m = 0;m < C; m++) { x[i] = a[k][m]; k = (k + 1) % R; i = i + 1;} }

Referring to FIGS. 4 and 5A, modulation module 420 comprises an adaptivetone mapping module 550, preamble insert module 555, a frequency-domainpre-emphasis filtering module 560, an adaptive notching module 565, aninverse fast fourier transform (IFFT) module 570 and a cyclic prefixmodule 575. Herein, adaptive tone mapping module 550 receives packets ofdata from interleaver 530 and uses PSK modulation to map bits of data toa corresponding analog frequency.

Thereafter, according to one embodiment, preamble insertion module 555is responsible for inserting a preamble into the start of each PHYframe. The preamble is used in transmissions to identify that the framehas arrived and the type of modulation conducted on the data within theframe.

It is contemplated that every time a signal is transmitted over a powerline, the signal will experience some attenuation, especially at thehigher frequencies. Hence, in order to compensate for attenuation causedby channel characteristics, frequency-domain pre-emphasis filter module560 is adapted to multiply complex elements received from adaptive tonemapping module 550 (e.g., complex frequency domain samples of an OFDMsymbol) with corresponding filter coefficient values. Each of the filtercoefficient values represents attenuation for one of the tones rangingfrom 0 dB up to 12 dB or higher, and such filter coefficient values canbe dynamic or static in nature.

More specifically, frequency-domain pre-emphasis filter module 560 mayinclude a multiplexer that is adapted to multiply the complex frequencydomain samples of an OFDM symbol with 128 real filter coefficients, thenconduct four (4) right shifts at the output. According to one embodimentof the invention, the filter coefficients are 5-bits representingunsigned values from 0 h to 10 h, and the filter coefficients is not beallowed to have values larger than 10 h. Of course, in otherembodiments, other bit sizes may be used. For this embodiment, thefilter multiplies the first 128 frequency-domain complex samples of anOFDM symbol with the 128 real coefficients of the filter. The rest ofthe 128 frequency-domain samples of the OFDM symbol typically are set tozero and are not be multiplied by the filter coefficients. As shown inFIG. 5H, input complex samples may be 8 bits each while the filtercoefficients may be 5 unsigned bits each. Since the maximum allowedvalue of any filter coefficients may be 10h, the output of themultiplication may be 12 bits (not 13 bits). The output may then beright shifted by 4 to get a final output of 8 bits that may be used asinput to the IFFT.

The filter coefficient values may vary from 0 to 16, and since four (4)right shifts are performed at the output. Hence, frequency-domainpre-emphasis filter module 560 may provide the following attenuation forany of the 128 carriers:

Scaling factor attenuation in dB 16/16 0 dB 15/16 −0.53 dB 14/16 −1.16dB 13/16 −1.8 dB 12/16 −2.5 dB 11/16 −3.25 dB 10/16 −4 dB  9/16 −5 dB 8/16 −6 dB  7/16 −7.2 dB  6/16 −8.5 dB  5/16 −10.1 dB  4/16 −12 dB 3/16 −14.5 dB  2/16 −18 dB  1/16 −24 dB  0/16 −infinite

For instance, as an illustrative example shown in FIG. 7, an OFDM signal700 propagating through the power line may be attenuated by as much as12 dB during transmission through the power line. As a result, thefrequency-domain pre-emphasis filter adjusts for such attenuation sothat the attenuated portions of OFDM signal 700 are adjusted for the 12dB loss. As a result, for this example, the higher tone frequencies 710associated with OFDM signal 700 is attenuated to be two times larger inthan lower tone frequencies 720 of signal 700. The compensation isnormally static in nature based on prior testing, although it may bedynamic in nature or programmable.

Referring back to FIG. 5A, adaptive notching module 565 is configured toutilize data from frequency-domain pre-emphasis filter module 560 inorder to place programmable notches in the frequency spectrum withoutany extra filtering. Hence, the adaptive notching scheme enablestransmitter 400 to avoid data transmission over selected frequencies andmay enable compliance with regional spectral/interference regulations.This notching may occur at the ends of the frequency spectrum, or mayoccur at selected frequencies within the frequency spectrum itself.

For instance, adaptive notching module 565 operates as a multiplier to“zero out” tones within the N-complex elements of the IFFT input vector.Based on a user-defined table, microcontroller 320 located within OFDMPLC PHY circuitry 300 of FIG. 3 is adapted to control the operations ofadaptive notching module 565 of FIG. 5A by altering the filtercoefficient values. This creates a different mapping table that isstored into registers accessed by the adaptive tone mapping module 550.The new mapping table will exclude certain frequency regions within aselected frequency band.

For instance, as an illustrative example, the CELENEC A standardsupports a frequency spectrum ranging between 9 kHz and 95 kHz at asample rate of 1.2 mega-samples. In order to notch all frequencies below20 kHz, adaptive notching module 565 computes the kilohertz range foreach IFFT complex element (N). For instance, where N is equal to 256,each IFFT complex element corresponds to 4.688 kHz. Hence, in order toavoid frequencies below 20 kHz, adapter tone mapping module 550 wouldneed to set filter coefficient values for the frequencies associatedwith the first four (4) IFFT complex elements to zero, and thus, notchthe frequency region between DC and 20 kHz.

However, in the event that the notch is to occur within the frequencyspectrum, there may be overlapping between tones and thus it may requirethe number of IFFT complex elements to be expanded for this calculation.This would cause the “N” value, constituting the number of IFFT complexelements, to be multiplied by a selected multiplier M in order toincrease resolution. As an example, where M is equal to four (M=4), thisexpands IFFT in order to notch frequency 50 kHz.

After frequency-domain pre-emphasis filter module 560 has performed thenecessary attenuation offsets and adaptive notching module 565 hasperformed any desired frequency notching operations, IFFT module 570using a N-bit IFFT input vector to formulate time domain OFDM words thatwill precede the cyclic prefix. In other words, IFFT module 570 convertsinformation in the frequency domain into information in the time domainbecause subsequent operations on the channels are performed in the timedomain.

Thereafter, cyclic prefix module 575 is responsible for mitigating ISIby creating a gap between the transmitted symbols. In other words, theprogrammable cyclic prefix length can be modified according to the delayspread of the channel. The reason is that the channel creates an anomalyby spreading the tones and the tone of one symbol too closely boarders atone of another symbol. By adding the gap, this could prevent ISIinterference.

Referring to FIGS. 4 and 5A, spectral shaping module 430 comprises an RCshaping module 580 and a time-domain pre-emphasis filter 585. RC shapingmodule 580 smoothes the OFDM signal by removing high frequencycomponents. Time-domain pre-emphasis filter 585 provides a linearequalization method where the transmit signal spectrum is shaped tocompensate for amplitude distortion and to further compensate thetransmit signal for attenuation introduced to the signal as itpropagates through the power line in the event that frequency domainpre-emphasis filter 560 is not able to accomplish the appropriateattenuation. As a result, time-domain pre-emphasis filter 585 operatesas a shelving filter that performs both low pass and high passfiltering, where the gain may be adjusted during the low pass filteringin order to adjust the attenuation of the signal at the higherfrequencies.

Referring back to FIG. 4, exemplary embodiment of an OFDM receiver 450supporting multi-tone OFDM based communications within a prescribed lowfrequency region is shown. Receiver 450 comprises a plurality of logicmodules including a data recovery module 460, demodulation module 470, adecoding module 480 and a channel reconfiguration module 490. Blindchannel quality measurement scheme is used to adaptively change thecoding and modulation used in a link in order to achieve reliablecommunications.

Referring now to FIG. 6, data recovery module 460 of OFDM receiver 450comprises a programmable gain amplifier (PGA) module 610, a channelequalizer module 615, DC blocker and jammer cancellation modules 620, aroot-mean-squared (RMS) module 625, an automatic gain control (AGC)module 630, a synchronization detection module 635 and a fast fouriertransform (FFT) module 640.

First, PGA module 610 is adapted to manage the amplification of anincoming signal arriving at receiver 450 via power line 220. PGA module610 measures the incoming signal and determines whether the signal isweak (gain below a predetermined threshold). If so, PGA module 610adjusts the gain of amplifier 615 in order to amplify the incomingsignal in order to achieve a sufficient amplification level fordemodulation.

The adjusted signal is routed to DC blocker and jammer cancellationmodules 620. DC blocker module 621 is adapted to remove the DC offsetfrom the incoming signal since analog-to-digital converters and analogfront-end circuitry are expected to apply at least some DC residual.Jammer cancellation module 622 is configured to detect the presence of ajammer signal and to remove it from the incoming signal. The removal ofthe DC offset and jamming signals is designed to avoid an incorrectreading by AGC module 630, which is designed to normalize the inputsignal to a predetermined power level. The presence of interference onthe signal may affect gain adjustment of the incoming signal.

RMS module 625 is designed to measure the power of the signal and/or thesignal-to-noise ratio (SNR) for use by channel estimation module 655 asdescribed below.

After being adjusted by AGC module 630, which may be adapted to tracksignal variations and normalizes the received signal to be the properbit size in order to ensure that soft Viterbi decoder 675 (below)operates optimally, the incoming signal is provided to synchronizationdetection module 635, which analyzes the contents of the incoming signalto detect the beginning of the preamble symbols and the data symbols.Moreover, synchronization detection module 635 determines the mode ofoperation of the current PHY frame (Normal or Robust).

As soon as the start of the data symbol is determined and the channel,such data is routed to FFT module 640 to convert the time domaininformation into the frequency domain. Thereafter, a channel equalizermodule 645 performs an adaptive frequency-domain channel equalization(FEQ) technique that compensates for severe attenuation in the channelat high frequencies by equalizing the received signal based on thereceived signal level of each tone.

As an example, channel equalizer module 645 may feature a tableincluding a set of complex coefficients that are used to correct theamplitude and phase of each sub-carrier where OFDM is used formodulation. The FEQ technique performed by module 645 corrects for thephase and the amplitude distortion introduced into each subcarrier bymultiplying the received constellation point of a subcarrier (in thefrequency domain) with a complex value that is adaptive (e.g., computeusing a least mean squares update algorithm). By being made adaptive,channel equalizer module 645 will be able to adjust for variationsbetween various channels, and also, will be able to track variations inthe same channel that may occur due to temperature variations and otherparameters. In other words, channel equalizer module 645 is adapted toaddress frequency selective, fading channels. Of course, channelequalizer module 645 could be static in an alternative embodiment.

The output data from channel equalizer module 645 is provided todemodulator module 470. As shown in FIGS. 4 and 6, demodulator module470 comprises a demodulator 650 that demodulates the incoming signal inorder to recover data contained therein. As an illustrative embodiment,the incoming signal is demodulated in accordance with a selecteddemodulation scheme such as differential binary phase shift keen(DBPSK). The output from demodulator 650 is provided to channelreconfiguration module 490.

Channel reconfiguration module 490 comprises channel estimation module655 that utilizes the SNR measurements as measured by RMS module 625 todetermine whether the signal quality is acceptable to operate a Normalmode or whether receiver 450 needs to operate in ROBO mode. In general,channel estimation module 655 performs two operations.

First, channel estimation module 655 determines whether the incomingsignal exceeds a predetermined SNR value. If the incoming signalfeatures signal-to-noise (SNR) ratio that exceeds the predetermined SNRvalue, receiver 450 will enter into Normal mode and will signal itstransmitter via the MAC layer to transmit a message for receipt by thedevice featuring transmitter 400 to enter into ROBO mode as well.However, if the SNR levels are satisfactory but the channel is adverselyaffecting the characteristics of a signal, such as not sufficientlydetecting the tones or grossly attenuating the tones, receiver 450 willenter into ROBO mode and signal the corresponding transmitter to enterinto ROBO mode.

Besides channel estimation, receiver 450 may signal the correspondingtransmitter 400 to adjust the number of tones in the event that thechannel attenuation eliminates certain tones and the elimination of suchtones does not adversely affect the operations of the system. The tonenumbering is set by a tone setting module 660 implemented within channelreconfiguration module 490.

Based on whether receiver 450 is operating Normal mode or ROBO mode,de-interleaver module 665 of decoding module 480 recovers the datadistributed by the interleaver module of transmitter 400 and providessuch data to a soft robust decoder module 670 or directly toconvolutional decoder module 675, a Reed-Solomon decoder module 680 anda descrambler module 685 in order to recover the data.

It is noted that AGC module 630 improves the performance of the softrobust decoder module because by normalizing the input data ensure thatsoft robust decoder module 670 receives enough bits to predict the rightinformation. Herein, soft robust decoder module 670 is in communicationwith de-interleaver module 665 that makes a determination as to the bitvalue based on repetitive bits that it analyzes. In other words, softrobust decoder module 670 is an intelligent component that makes adecision as to what is the right value associated with the repetitivedata bits in order to fully utilize the coding gained from Robust mode.This is accomplished by giving more weight to sub-carriers of betterquality and less noise (e.g. based on SNR measurements) because, morelikely than not, the data of these sub-carriers is accurate.

While the invention has been described in terms of several embodiments,the invention should not be limited to only those embodiments described,but can be practiced with modification and alteration within the spiritand scope of the invention as will be claimed.

What is claimed is:
 1. A first network device connected by a power lineto a distribution transformer, the first network device comprising: atransceiver to communicate with a second network device via the powerline through the distribution transformer in a frequency band from 1 kHzto 600 kHz; a tone mapping module to select orthogonal tones in thefrequency band for modulation based on a mapping table; and apre-emphasis module to pre-emphasize at least one of the selectedorthogonal tones to compensate for attenuation of a signal transmittedthrough the power line, wherein the transceiver receives a channelestimate generated by the second network device based on the signal toadjust the mapping table and pre-emphasis.
 2. The first network deviceof claim 1, wherein the tone mapping module and the pre-emphasis modulerespectively adjust the mapping table and pre-emphasis based on thechannel estimation received from the second network device.
 3. The firstnetwork device of claim 1, further comprising an interleaver tointerleave data, to operate in a plurality of modes, and to performrepetitive encoding on the data in one of the plurality of modes.
 4. Thefirst network device of claim 3, further comprising an encoder torandomize the data and to insert error correction data.
 5. The firstnetwork device of claim 3, further comprising a modulator to modulatethe data to the selected orthogonal tones.
 6. The first network deviceof claim 3, further comprising a selector to select one of the pluralityof modes of the interleaver based on the channel estimate.
 7. The firstnetwork device of claim 3, wherein: the interleaver randomizes the dataduring a first one of the plurality of modes, and reduces a data rateand performs the repetitive encoding in a second one of the plurality ofmodes; and the repetitive encoding includes repeating bits and placingeach repetition of the bits at different frequencies and withindifferent symbols.
 8. The first network device of claim 3, wherein: theinterleaver comprises a plurality of sub-interleavers; and at least oneof the sub-interleavers reverses an order of packets.
 9. The firstnetwork device of claim 1, further comprising a preamble insertionmodule to insert a preamble into each frame to identify a type ofmodulation performed on data within the frame.
 10. The first networkdevice of claim 1, further comprising a filter module to multiply atleast one of the selected orthogonal tones by a coefficient tocompensate for attenuation caused by channel characteristics.
 11. Thefirst network device of claim 1, further comprising a notching module toplace programmable notches in the frequency band to avoid datatransmission over the one or more of the selected orthogonal tones. 12.The first network device of claim 1, wherein the power line is analternating current power line.